A 16 KBS Full Duplex
Spread Spectrum Receiver RF Data Link
By: Dan Doberstein,
President DKD Instruments
Preface to this Reprint
This article first appeared in SSS(Spread SPectrum Scene)
in Oct 1992. In addition to the figures presented in that article
a number of others have been included from my notes at that time.
I hope that the reader finds the article interesting. I never had
the time to finish all parts of this receiver but look forward to
some volunteer help. It would be an excellent senior project for a
interested engineering student. I offer my help for any interested.
I can also provide IC's and components.
Dan Doberstein, Aug 1996
Part I, A 16KBS Spread Spectrum Full Duplex RF Data
Link
This article describes a full duplex Direct Sequence RF data link
using the cordless telephone chip set. Although this frequency band
is not allocated for spread spectrum use it is legal at low power
levels. With a couple of mixers and associated LO's it can be translated
to other bands.
The system block diagram is shown in figure 1. The MC145168 and the
MC3362 form the heart of the system. The MC14568 is a dual PLL that
locks both the xmiter and receive LO's. The frequencies used for xmitt
and receive are burnt into the internal ROM of the MC145168, in other
words not change-able unless the reference frequency is altered. A
table of 16 xmitt and 16 receive frequency pairs are available in
the 45 to 49 mhz band.The channel select inputs are used to select
a xmit/rec pair.
The MC3362 is a single chip dual conversion receiver that was designed
for the cellular and cordless telephones. In this design it is used
to convert the received signal to a first IF of 10.7 Mhz. and then
to the 2nd IF of 455 KHz. A single tuned bandpass is used as the first
IF filter. A bandwidth of about 1 mhz is needed here as the signal
is still wide band at this stage, ie. rec' signal is not despread
at this point. The 1 mhz BW is a Q of about 10 in the 10.7 filter,
probably best obtained with a simple LC filter. The 1 Mhz BWcomes
from the chip rate of approx 1 MHZ.
The 10.7 is mixed with the 10.24 MHz LO which has thespreading code
impressed on it. The LO is phase modulated with the replica of the
xmitt code.Assuming correlation of the receiver generated code(ie
LOCK) the output of the 10.7 mixer is a despread carrier at 455 khz.
This carrier will still have our biphase modulated data on it. The
filter following the 455 IF is a bandpass filter with BW equal to
about 12 KHz(Toko Ceramic). This is just enough to pass our data modulation
stream of 16KHz. The data stream can be user data or 16KBS datafrom
the CVSD audio coder/decoder chip , MC3418. The data is biphase modulated
on the carrierusing exclusive OR gates. In addition to the data the
spreading code is also EX-OR'ed to this datastream.
The resultant data stream then is applied to biphase
modulator at xmitter. The MC3418 chip is a Continuously Variable Slope
Demod/Mod . It takes speech or other analog signals and converts them
to a lower bit rate serial data stream than one would get using straight
ahead sampling theory, ie twice the highest freq component for sample
rate. This would be about 10Khz for speech and at 8 bits/sample that
works out to a 80 KBS serial data stream. The CVSD substantially reduces
the data rate, some quality of reproduced signal is lost though. The
chip not only provides the modulation but also the demodulation.
One of the design goals of this system was to simplify the frequency
plan. This has cost as well as practical benefits. The PLL's, code
clocks, and data clocks all are locked to or derived from one xtal
oscillator at 10.24 MHz. Another advantage is that with clock coherence
the data clock is automatically recovered once code lock is achieved.
This simplifies data recovery problem considerably.
There are two code generators for each station, one for xmitt and
one for receive. The receive code generator is used to correlate with
the incoming signal. The 10.24 Mhz clock is divided by 10 to obtain
the xmitt code clock. The rec' code generator is clocked by the Code
Sync and Clock Generator circuitry. The code used for xmitt is WILL
NOT be the same as for received. One code pair will be used for each
station pair. The user will chose which code pair used by changing
jumpers on code generators. The use of differennt xmit/rec codes helps
with rejection of xmitt signal into recieve channel due to low correlation
of these two signals, ie xmitt signal is spread, NOT despread, in
rec channel.
The code Sync and Clock Generator circuitry provides a modulated version
of the 10.24 MHZ. reference divided by 10 to give approximately 1.024
Mhz code clock for rec' code generation. The 1.024 MHz clock is modulated
in such a way as to keep the receiver generated replica of the xmitters
code aligned with incoming code on received signal.At the heart of
the code lock circuit is the tua dither method of code tracking. The
details are dicussed in the next section of this article.
Code Lock Circuitry
This section explains the code lock circuitry. The code lock systems
job is to keep the receivers code locked or correlated with the transmitted
code. The technique used is a modified Tau Dither system. Referring
to the block diagram we see the 10.24 Mhz clock is modulated by the
divide by 9/10/11 circuitry before it is passed to the code generator.
The divide by 9/10/11 block serves the same purpose as a VCXO code
clock in a conventional Tau Dither circuit.
The divide by 9/10/11 provides the mechanism to modulate the code
clock so as to keep the receivers code locked to the transmitters.
This circuit sends most of the time in the divide by 10 state. When
a rate clock pulse is detected the 10.24 Mhz clock is divided by 9/11
for one cycle. The net effect is that for every rate pulse one input
clock pulse is added or subtracted depending on the 9/11 input. The
rate control input functions as a gain control point. The higher the
rate clock frequency the higher the gain for a given ADV/RET command.
Lower rate clock frequencies result in slower reaction to ADV/RET
commands. Jitter during code lock will always be +/- 1/10 of a chip
because of the discrete nature of the clock modulation.
Modulated code clock is sent to the code generator. The output of
the code generator is split with one path passing through a delay
element. The dithered code is created by toggling between the delayed
and undelayed versions of the code. The rate of toggling is set by
the dither clock.
The dithered code is used to phase modulate the 10.24 Mhz clock which
here is used as the 2nd LO. The modulated LO is then bandpassed to
bandlimit the resultant wide band signal. The LO is now mixed with
the received signal which creates the 455 Khz IF. The spectrum shown
for the 455 IF assumes code lock. If the codes are not locked the
455 IF would just be noise. The despread 455 IF still has the data
modulation on it so its bandwidth is about 16Khz.
In addition to the data phase modulation on the 455 IF there is Amplitude
Modulation induced by the dithering process. The dither induced AM
is itself phase modulated via the correlation process. In order to
keep the receivers code locked this AM signal must be recovered and
processed to provide the error signal used to drive code misalignment
to zero (+/- 1/10 chip). After bandpassing and detection we now have
the desired AM component, the dither signal. A bandpass centered at
the dither frequency is used to separate the dither AM signal from
other AM signals which may be present on the 455 IF. The output of
the band pass is hardlimited via the zero crosser and passed to the
Dither Phase Detector. This process ignores the amplitude information
contained in the dither signal and concerns itself only with the PHASE
information. This is a simplification of the text book Tau Dither
technique where both the phase and amplitude of the Dither signal
are used. This is possible since the phase modulation on the dither
signal contains the code advance/retard information while the amplitude
of the dither signal contains the "how much" information.
In short the circuit disregards the amplitude information present
in the dither signal and uses just the phase information to maintain
code lock. This simplification has a price in that higher SNR's are
needed to maintain and obtain code lock.
It is easy to get confused here with all the modulations present.
Remember the 455 Khz IF has the 16Kbs Biphase modulation on it and
the dither induced AM is itself Biphase modulated. These two Biphase
modulations are separate and distinct from each other and can be processed
independently from each other as done here.
The phase of the dither signal is recovered by using an EXOR gate
and comparing with the dither reference clock. The signal is low passed
using the loop filter. The filter serves as an averager. The This
filter in large part determines Lock range, Pull In range, Steady
State code alignment error and the general dynamic behavior of the
closed loop code tracking process. After zero crossing detection the
signal is passed to the divide by 9/10/11 circuit. The signal out
of the loop filter is also a measure of code clock frequency offsets
between the transmitters code clock and the receivers code clock.
If the offset is zero, i.e the code clocks are exactly the same frequency,
the average value of the error will be exactly zero, or equivalent
zero bias DC value. If the frequencies are not equal, the usual case,
the error signal will have a non zero average value. Depending on
the circuit you could run out of "headroom", hit your voltage
rails, and break lock. This imposes a limit on the amount of code
clock offset allowed between transmitter and receiver code clocks.
It should be noted that the high dynamics from excessive frequency
offsets can itself lead to failure to obtain or maintain lock.
We have closed the code tracking loop and only have the TRACK/SCAN
switch left to explain. this switch is controlled by the carrier detect
output of the MC3362. When no carrier is present the switch is set
to scan which holds the 9/10/11 circuit in the RETARD position for
scanning purposes. This switch ensures the searching in one direction
as without it a random search caused by noise on ADV/RET control line
would result.
Thats it for this month, next month we will discuss the CVSD Data
Mod./Demod. and the Costas Loop Demodulator.
16KBS Full Duplex S.S. RF Data Link , Part III
This section examines the data demodulator and the audio reconstruction
using the CVSD demodulator.
A Costas Loop demodulator is used to recover the data from the 455
Khz IF. Of course we must have code lock before an IF would be present.
Assuming code lock the 455 KHz carrier has just the data modulation
on it, the remaining dither modulation being negligible. The incoming
carrier is first limited to remove any amplitude variation. Since
the information is carried in the PHASE of the carrier this operation
does not destroy any data information. After limiting it is split
and sent to two identical phase detectors. Classic Costas loop demodulators
use three, four quadrant, multipliers. This design approximates these
multipliers with two phase detectors and one double balanced modulator.
The Exar part XR2211 contains two phase detectors, a VCO with 0 and
90 degree ouputs, Input limiter and the Inphase limiter. The MC1496
does the chopper/modulator function. The two phase detectors compare
the phase of the carrier with the VCO output, one at 0 degrees, the
other at 90 degrees. Two channels are now present, the I or Inphase,
and the Q or Quadrature. The Inphase channel carries the recovered
data as shown in the timing diagrams.
The output of the phase detectors are lowpassed with the bandwidth
approximately equal to the data rate. The inphase channel is now limited
and sent to the MC1496. This limited signal is used to switch the
output of the MC1496 from invert to noninvert state. Hence the chopper
designation. To achieve this the MC1496 is used in its Double sideband,
Suppressed carrier mode. Essentially the Inphase channel affects only
the SIGN of the MC1496 output not its magnitude. The MC1496 output
is then lowpased by the Loop filter and passed to the VCO control
point. The loop filter serves exactly the same function as the loop
filter in a standard PLL. The Loop, Inphase and Quadrature filters
are all single pole RC types. The Loop filter should have a time constant
of about 2 to 10 msec.
The timing waveforms show operation assuming the loop has acquired
the carrier, in other words the frequency error between the VCO and
the IF is zero and the phase error is small. If there is any frequency
offset between the free running VCO frequency and the IF (there always
is!) a small DC bias level will exist at the output of the Loop filter
to correct out this constant frequency error. If this offset is to
large, or changes with time i.e doppler shift caused by excessive
receiver or transmitter relative movement, carrier acquisition will
not be obtained or ,in the changing case, maintained. This "bias"
is illustrated in waveform 6.
The CVSD demodulator is fed from the resynced data and reconstructs
the audio signal from the serial bit stream. The 16KHz clock is only
valid when the we have code lock condition. Without code lock the
16Khz clock will not be synchronous with the transmitted data clock.
One of the primary advantages to coherent Direct sequence systems
is that data clock synchronism is achieved simultaneously with code
lock. The MC3417 Continuously Variable Slope Demodulator/Modulator
converts the audio signal efficiently to and from a low bit rate serial
data streams. It also does another important job by insuring a changing
bit stream during "quiet" periods. Excessively long strings
of all ones or zeros can create problems in receiver operation. Next
month we finish up this design article with some thoughts on microprocessor
interface for code selection ,a serial data interface and using the
Fujitsu dual PLL MB1519 instead of the Motorola MC145168.
16KBS Spread Spec full duplex design
Part 4
This section dicusses a possible solution for code generation,
alternative PLL and CVSD solutions. Figure XX shows the code generator
block diagram. Instead of the tried and true shift register approach
a 4K x 1bit piece of static RAM is used. A counter/address generator
is driven by the dithered code clock which in turn generates sequential
addresses which clock the code out of the RAM. This architecture allows
complete code flexibility since codes can be generated by the controlling
computer and downloaded to the RAM. A start address and code length
word are also sent. This allows storing multiple codes in the RAM
and accessing them on the fly.
Switching codes on the fly could be used to implement a two code system
where acquisition is done using a short code of say 15 bits and after
code lock switch to a much longer code. Longer codes increase the
number of users (within limits) per channel and decrease the probability
that another user will be "on" your code. The down side
of longer codes is increased acquisition time especially with the
sliding correlation method used here, hence the short/long code solution.
It is the synchronizing of the switch between the codes that is tough!
One method would be to deliberately put a small offset (in code bits)
in the receiver long code wrt to the transmitted code at the moment
when the codes are switched. This will temporally break lock but by
forcing the receiver to search in one direction only,towards lock,
you will quickly achieve lock again on the long code.
An interesting replacement for the MC3417 CVSD is the
Harris CVSD chip set. The HC55536 does the demodulation operation
while the HC55564 does the modulation operation. Operation from 9KBS
to over 64KBS is claimed. The design replaces the analog filters used
in the Motorola design with internal digital ones. This results in
fewer components for a complete solution.
Another more flexible choice for the PLL is the Fujitsu MB1519. This
dual PLL goes to 600 Mhz and has a user programmable divider using
swallow A counter /N counter technique. A three wire serial interface
is used so minimum hardware connection to computer is needed. The
reference divider has only two values, 512 and 1024. This forces channel
spacing of 20 Khz or 10 Khz with a 10.24 MHZ reference. By adding
a divide by 2 prescaler, NEC 584, we can double our VCO max frequency
to 1200 MHz. Of course at these higher frequencies we are losing image
rejection with that 1st IF of 10.7 Mhz, so another IF in the receiver
chain may need to be added.
Well that's it for this article, I hope you are inspired to try some
of this out and at the least found it interesting reading.
List of Figures
figure 1
figure 2
figure 3
figure 4
figure 5, Code Clock Shift Circuit
figure 6
figure 7, Possible PAL Implementation
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